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HV9911 Switch-Mode LED Driver IC with High Current Accuracy
Features
Switch mode controller for single switch drivers o Buck o Boost o Buck-boost o SEPIC Works with high side current sensing Closed loop control of output current High PWM dimming ratio Internal 250V linear regulator (can be extended using external zener diodes) Internal 2% Voltage Reference (0C < TA < 85C) Constant frequency or constant off-time operation Programmable slope compensation Enable & PWM dimming +0.2A/-0.4A gate drive Output short circuit protection Output over voltage protection Synchronization capability Programmable MOSFET current limit Soft start
General Description
The HV9911 is a current mode control LED driver IC designed to control single switch PWM converters (buck, boost, buck-boost, or SEPIC), in a constant frequency or constant off-time mode. The controller uses a peak current control scheme, (with programmable slope compensation), and includes an internal transconductance amplifier to control the output current in closed loop, enabling high output current accuracy. In the constant frequency mode, multiple HV9911s can be synchronized to each other, or to an external clock, using the SYNC pin. Programmable MOSFET current limit enables current limiting during input under voltage and output overload conditions. The IC also includes a 0.2A source and 0.4A sink gate driver for high power applications. An internal 9 to 250V linear regulator powers the IC, eliminating the need for a separate power supply for the IC. HV9911 provides a TTL compatible, PWM dimming input that can accept an external control signal with a duty ratio of 0-100% and a frequency of up to a few kilohertz. The IC also provides a FAULT output which, can be used to disconnect the LEDs in case of a fault condition, using an external disconnect FET. The HV9911 based LED driver is ideal for RGB backlight applications with DC inputs. The HV9911 based LED lamp drivers can achieve efficiency in excess of 90% for buck and boost applications.
Applications
RGB backlight applications Automotive LED driver application Battery Powered LED lamps Other DC/DC LED drivers
Typical Application Circuit - Boost
CIN
1 VIN GATE 3
L1
D1
Q1
ROVP1
CO
CDD
2 VDD CS 5
RSC
RCS
ROVP2
4
GND
OVP
12
RSLOPE
6 SC
HV9911
FAULT 11
Q2
RT
7
RT
FDBK
16
CREF
Cc
10 REF COMP 14
Rs
RL2
9
CLIM
PWMD 13
RL1
15 IREF SYNC 8
RR1 RR2
HV9911 Typical Application Circuit - Buck
Rs HV7800
CIN
1 VIN OVP 12
CO D1
CDD
2 VDD FAULT 11
L1
4
GND
GATE
3
Q1
RSLOPE
6 SC
HV9911
CS 5
RT
7
RSC
RT COMP 14
RCS
CREF
Cc
10 REF
FDBK 16
RL2
9
CLIM
PWMD 13
RL1
15 IREF SYNC 8
RR1 RR2
Typical Application Circuit - SEPIC
L1 C1 CIN
1 VIN GATE 3
D1
Q1
ROVP1
CO
L2 RSC RCS ROVP2
CDD
2 VDD CS 5
4
GND
OVP
12
RSLOPE
6 SC
HV9911
FAULT 11
Q2
RT
7
RT
FDBK
16
CREF
Cc
10 REF COMP 14
Rs
RL2
9
CLIM
PWMD 13
RL1
15 IREF SYNC 8
RR1 RR2
2
HV9911 Ordering Information
DEVICE HV9911
-G indicates package is RoHS compliant (`Green')
Pin Configuration
Package Options 16-Lead SOIC HV9911NG-G
1 2 3 4 5 6 7 8
VIN FDBK
16 15 14 13 12 11 10 9
VDD
IREF
GATE
COMP
GND
PWMD
HV9911
CS OVP FAULT SC
RT
REF
SYNC
CLIM
Absolute Maximum Ratings
Parameter VIN to GND VDD to GND CS1, CS2 to GND PWMD to GND GATE to GND All other pins to GND Continuous Power Dissipation (TA = +25C) 16-Pin SOIC (derate 10.0mW/C above +25C) Junction to ambient thermal impedance Operating ambient temperature range Junction temperature Storage temperature range 1000mW 82OC/W -40C to +85C +125C -65C to +150C Value -0.5V to +250V -0.3V to +13.5V -0.3V to (VDD + 0.3V) -0.3V to (VDD + 0.3V) -0.3V to (VDD + 0.3V) -0.3V to (VDD + 0.3V)
Stresses beyond those listed under "Absolute Maximum Ratings" may cause permanent damage to the device. These are stress ratings only, and functional operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to absolute maximum rating conditions for extended periods may affect device reliability.
3
HV9911 Electrical Characteristics
(Over recommended operating conditions. VIN = 24V, TA = 25C, unless otherwise specified)
Symbol Input VINDC IINSD
Parameter
Min
Typ
Max
Units
Conditions
Input DC supply voltage range* Shut-down mode supply current*
(1)
1.0
250 1.5
V mA
DC input voltage PWMD connected to GND, VIN = 24V
-
Internal Regulator VDD UVLO UVLO VDD(ext) Internally regulated voltage* VDD undervoltage lockout threshold VDD undervoltage lockout hysteresis Steady state external voltage that can be applied at the VDD pin2 7.25 6.65 7.75 6.90 500 8.25 7.20 12 V V mV V VIN = 9-250V, IDD(ext) = 0, PWMD connected to GND VDD rising -----
Reference REF pin voltage (0C < TA < 25C) REF pin voltage (-40C < TA < 85C) Line regulation of reference voltage Load regulation of reference voltage 1.225 1.2125 0 1.25 1.25 1.275 V 1.275 20 mV REF bypassed with a 0.1F capacitor to GND; IREF= 0; VDD = 7.75V; PWMD = GND REF bypassed with a 0.1F capacitor to GND; IREF = 0; VDD = 7.25 - 12V; PWMD = GND REF bypassed with a 0.1F capacitor to GND; IREF = 0-500; PWMD = GND VDD = 7.25V - 12V VDD = 7.25V - 12V VPWMD = 5.0V VGATE = 0V; VDD = 7.75V VGATE = 7.75V ; VDD = 7.75V CGATE = 1nF; VDD = 7.75V CGATE = 1nF; VDD = 7.75V
VREF
VREFLINE VREFLOAD
0
-
10
mV
PWM Dimming VPWMD(lo) VPWMD(hi) RPWMD GATE ISOURCE ISINK TRISE TFALL GATE short circuit current GATE sinking current GATE output rise time GATE output fall time 0.2 0.4 50 25
-
PWMD input low voltage* PWMD input high voltage* PWMD pull-down resistance
2.0 50
100
0.80 150
V V k
A A ns ns
85 45
Over Voltage Protection VOVP IC shut down voltage* 1.215 1.25 1.285 V VDD = 7.25 - 12V ; OVP rising
Current Sense TBLANK TDELAY1 Leading edge blanking Delay to output of COMP comparator 100 375 180 ns ns --COMP = VDD ; CLIM = REF; VCS = 0 to 600mV step
4
HV9911
Symbol
TDELAY2
Parameter
Delay to output of CLIMIT comparator
Min
-
Typ
-
Max
180
Units
ns
Conditions
COMP = VDD ; CLIM = 300mV ; VCS = 0 to 400mV step
VOFFSET GB AV VCM VO gm VOFFSET IBIAS Oscillator fOSC1 fOSC2 DMAX IOUTSYNC IINSYNC
Comparator offset voltage
-10
-
10
mV
---
Internal Transconductance Opamp Gain bandwidth product# Open loop DC gain Input common-mode range Output voltage range# Transconductance Input offset voltage Input bias current
# #
66 -0.3 0.7 340 -2.0 -
1.0 435 0.5
3.0 6.75 530 4.0 1.0
MHz dB V A/V mV nA
75pF capacitance at COMP pin Output Open --VDD = 7.75V -------
Oscillator frequency* Oscillator frequency* Maximum duty cycle Sync output current Sync input current
88 308 0
100 350 90 10 -
112 392 20 200
kHz kHz % A A
RT = 909k RT = 261k ----VSYNC < 0.1V
Output Short Circuit TOFF TRISE,FAULT TFALL,FAULT GFAULT Soft Start ICLIM Current into CLIM pin when pulled low 200 A FAULT is low ; 06.25k resistor between REF and CLIM Propagation time for short circuit detection Fault output rise time Fault output fall time Amplifier gain at IREF pin 1.8 2 250 300 200 2.2 ns ns ns IREF = 200mV ; FDBK = 450mV; FAULT goes from high to low 1nF capacitor at FAULT pin 1nF capacitor at FAULT pin IREF = 200mV
Slope Compensation ISLOPE GSLOPE
1 2
Current sourced out of SC pin Internal current mirror ratio
0 1.8
2
100 2.2
A -
--ISLOPE = 50A ; RCSENSE = 1k
See application section for minimum input voltage. Parameters are not guaranteed to be within specifications if the external VDD voltage is greater than VDD(ext) or if VDD < 7.25V. * Specifications which apply over the full operating ambient temperature range of -40C < TA < +85C. # Guaranteed by design.
5
HV9911
Pin Description
Pin # 1 2 3 4 5 6 Pin VIN VDD GATE GND CS SC Description This pin is the input of a 250V high voltage regulator. This is a power supply pin for all internal circuits. It must be bypassed with a low ESR capacitor to GND (at least 0.1uF). This pin is the output gate driver for an external N-channel power MOSFET. Ground return for all circuits. This pin must be connected to the return path from the input. This pin is used to sense the drain current of the external power FET. It includes a built-in 100ns (min) blanking time. Slope compensation for current sense. A resistor between SC and GND will program the slope compensation. In case of constant off-time mode of operation, slope compensation is unnecessary and the pin can be left open. This pin sets the frequency or the off-time of the power circuit. A resistor between RT and GND will program the circuit in constant frequency mode. A resistor between RT and GATE will program the circuit in a constant off-time mode. This I/O pin may be connected to the SYNC pin of other HV9911 circuits and will cause the oscillators to lock to the highest frequency oscillator. This pin provides a programmable input current limit for the converter. The current limit can be set by using a resistor divider from the REF pin. Soft start can also be provided using this pin. This pin provides 2% accurate reference voltage. It must be bypassed with at least a 10nF - 0.22F capacitor to GND. This pin is pulled to ground when there is an output short circuit condition or output over voltage condition. This pin can be used to drive an external MOSFET in the case of boost converters to disconnect the load from the source. This pin provides the over voltage protection for the converter. When the voltage at this pin exceeds 1.25V, the gate output of the HV9911 is turned off and FAULT goes low. The IC will turn on when the power is recycled. When this pin is pulled to GND (or left open), switching of the HV9911 is disabled. When an external TTL high level is applied to it, switching will resume. Stable Closed loop control can be accomplished by connecting a compensation network between COMP and GND. The voltage at this pin sets the output current level. The current reference can be set using a resistor divider from the REF pin. This pin provides output current feedback to the HV9911 by using a current sense resistor.
7 8 9 10 11
RT SYNC CLIM REF FAULT
12 13 14 15 16
OVP PWMD COMP IREF FDBK
6
HV9911 Functional Block Diagram
VIN
Linear Regulator
Vbg
REF
VDD
POR GATE
CLIM
_
SS Blanking
100ns
+
FAULT
SC
+
SS
+
IREF COMP R One Shot
SS 2
Functional Description
Power Topology
The built in linear regulator of the HV9911 can operate up to 250V at the VIN pin. The linear regulator provides an internally regulated voltage of 7.75V (typ) at VDD if the input voltage is in the range of 9V - 250V. This voltage is used to power the IC and also provide the power to external circuits connected at the VDD and VREF pins. This linear regulator can be turned off by overdriving the VDD pin using an external boostrap circuit at voltages higher than 8.25V (up to 12V). In practice, the input voltage range of the IC is limited by the current drawn by the IC. Thus, it becomes important to determine the current drawn by the IC to find out the maximum and minimum operating voltages at the VIN pin. The main component of the current drawn by the IC is the current drawn by the switching FET driver at the GATE pin. To estimate this current, we need to know a few parameters of the FET being used in the design and the switching frequency. Note: The equations given below are approximations and are to be used only for estimation purposes. The actual values will differ somewhat from the computed values. Consider the case when the external FET is FDS3692 and the switching frequency is fS = 200kHz with an LED string voltage VO = 80V. From the datasheet of the FET, the following parameters can be determined: CISS = 746pF CGD = CRSS = 27pF CGS = CISS - CGD = 719pF VTH = 3V
7
_
FDBK
_
Gm
13R
PWMD
GND
+
Q
S
_
_
+
1:2
ramp S Q R POR Vbg OVP
+
CS
R
Q
_
SYNC RT
HV9911
CGD IPK
HV9911
RGATE
I1 Iavg 0 t1 t2 t3
VDD
CGS
Fig. 1. Current Sourced out of GATE at FET turn-on Driver The typical waveform of the current being sourced out of GATE is shown in Fig. 1. Fig. 2 shows the equivalent circuit of the gate driver and the external FET. The values of VDD and RGATE for the HV9911 are 7.75V and 40 ohms respectively.
Fig. 2. Equivalent Circuit of the Gate Driver When the external FET is being turned on, current is being sourced out of the GATE and that current is being drawn from the input. Thus, the average current drawn from VDD (and thus from VIN) needs to be computed. Without going into the details of the FET operation, the various values in the graph of Fig. 1 can be computed as follows.
Parameter IPK I1
Formula VDD RGATE VDD - VTH RGATE -RGATE * CISS * In I1 IPK
Value (for given example) 193.75mA
118.75mA
t1
14.61ns
( VO - VTH ) *CGD
t2 I1
(for a boost converter) 17.5ns (for a buck converter) 66ns 1.66mA
( VIN - VTH * CGD)
I1
t3 Iavg
2.3 * RGATE * CGS (I1 * [t1 + t2] + 0.5 * [IPK - I1] * t1 + 0.5 * I1 * t3) * fS
8
HV9911
The total current being drawn from the linear regulator for a typical HV9911 circuit can be computed as follows (the valCurrent Quiescent Current Current sourced out of REF pin ues provided are based on the continuous conduction mode boost design in the application note - AN-H55). Typical Value 1000A 100A
Formula 1000A VREF VREF + RL1 + RL 2 RR1 + RR 2 6V RT 1 2 .5 V * 2 RSLOPE 2 .5 V RSLOPE IAVG
Current sourced out of RT pin
13.25A
Current sourced out of SC pin
30.8A
Current sourced out of CS pin Current drawn by FET gate driver Total Current drawn from the linear regulator
61.6A 1660A 2.865mA
Note: For a discontinuous mode converter, the currents sourced out of the SC and CS pin will be zero.
Maximum Input Voltage at VIN pin computed using the Power Dissipation Limit
The maximum input voltage that the HV9911 can withstand without damage if the regulator is drawing about 2.8mA will depend on the ambient temperature. If we consider an ambient temperature of 40C, the power dissipation in the package cannot exceed PMAX = 1000mW - 10mW * (40OC - 25OC) = 850mW The above equation is based on package power dissipation limits as given in the Absolute Maximum Limits section of this datasheet. To dissipate a maximum power of 850mW in the package, the maximum input voltage cannot exceed P Vin max = max ITOTAL = 296 V Since the maximum voltage is far greater than the actual input voltage (24V), power dissipation will not be a problem for this design. For this design, at 24V input, the increase in the junction temperature of the IC (over the ambient) will be = VIN * ITOTAL * JA = 5.64OC where JA is the junction to ambient thermal impedance of the 16-pin SOIC package of the HV9911.
Minimum Input Voltage at VIN pin
The minimum input voltage at which the converter will start and stop depends on the minimum voltage drop required for the linear regulator. The internal linear regulator will regulate the voltage at the VDD pin when VIN is between 9 and 250V. However, when VIN is less than 9V, the converter will still function as long as VDD is greater than the under voltage lockout. Thus, the converter might be able to start at input voltages lower than 9V. The start/stop voltages at the VIN pin can be determined using the minimum voltage drop across the linear regulator as a function of the current drawn. This data is shown in Fig. 3 for different junction temperatures.
9
HV9911
14 13 12 11 25C -40C
Input Current (mA)
10 9 8 7 6 5 4 3 2 1 0 0.2 0.4 0.6 0.8 1 1.2 1.4 1.6 1.8 2
85C 125C
2.2
2.4
Minimum drop in Linear Regulator (V)
Fig. 3. Graph of input current vs minimum voltage drop across linear regulator for different junction temperatures Assume a maximum junction temperature of 85C (this give a reasonable temperature rise of 45C at an ambient temperature of 40C). At 2.86mA input current, the minimum voltage drop from Fig. 3 can be approximately estimated to be VDROP = 0.75V. However, before the IC starts switching the current drawn will be the total current minus the gate drive current. In this case, that current is IQ_TOTAL = 1.2mA. At this current level, the voltage drop is approximately VDROP1 = 0.4V. Thus, the start/stop VIN voltages can be computed to be: VINSTART = UVLOMAX + VDROP1 = 7.2V + 0.4V = 7.60V VINSTOP = UVLOMAX - 0.5V + VDROP = 7.2V - 0.5V + 0.75V = 7.45V Note: In some cases, if the gate drive draws too much current, VINSTART might be less than VINSTOP. In such cases, the control IC will oscillate between ON and OFF if the input voltage is between the start and stop voltages. In these circumstances, it is recommended that the input voltage be kept higher than VINSTOP. Note: In order to avoid abnormal startup conditions, the bypass capacitor at the REF pin should not exceed 0.22F.
Oscillator
The oscillator can be set in two ways. Connecting the oscillator resistor between the RT and GATE pins will program the off-time. Connecting the resistor between RT and GND will program the time period. In both cases, resistor RT sets the current, which charges an internal oscillator capacitor. The capacitor voltage ramps up linearly and when the voltage increases beyond the internal set voltage, a comparator triggers the SET input of the internal SR flip-flop. This starts the next switching cycle. The time period of the oscillator can be computed as: TS RT x 11pF
Slope Compensation
For converters operating in the constant frequency mode, slope compensation becomes necessary to ensure stability of the peak current mode controller, if the operating duty cycle is greater than 0.5. Choosing a slope compensation which is one half of the down slope of the inductor current ensures that the converter will be stable for all duty cycles. Slope compensation can be programmed by two resistors RSLOPE and RSC. Assuming a down slope of DS (A/s) for the inductor current, the slope compensation resistors can be computed as:
Reference
HV9911 includes a 2% accurate, 1.25V reference, which can be used as the reference for the output current as well as to set the switch current limit. This reference is also used internally to set the over voltage protection threshold. The reference is buffered so that it can deliver a maximum of 500A external current to drive the external circuitry. The reference should be bypassed with at least a 10nF low ESR capacitor.
Rslope =
10 x RSC DS x 10 6 x Ts x Rcs
A typical value for RSC is 499.
10
HV9911
Note: The maximum current that can be sourced out of the SC pin is limited to 100A. This limits the minimum value of the RSLOPE resistor to 25k. If the equation for slope compensation produces a value of RSLOPE less than this value, then RSC would have to be increased accordingly. It is recommended that RSLOPE be chosen in the range of 25k - 50k. pin which goes low during any fault condition. At startup, a monoshot circuit, (triggered by the POR circuit), resets an internal flip-flop which causes FAULT to go high, and remains high during normal operation. This also allows the gate drive to function normally. This pin can be used to drive an external disconnect switch (Q2 in the Typical Boost Application Circuit on pg.1), which will disconnect the load during a fault condition. This disconnect switch is very important in a boost converter, as turning off the switching FET (Q1) during an output short circuit condition will not remove the fault (Q1 is not in the path of the fault current). The disconnect switch will help to disconnect the shorted load from the input.
Current Sense
The current sense input of the HV9911 includes a built in 100ns (minimum) blanking time to prevent spurious turn off due to the initial current spike when the FET turns on. The HV9911 includes two high-speed comparators - one is used during normal operation and the other is used to limit the maximum input current during input under voltage or overload conditions. The IC includes an internal resistor divider network, which steps down the voltage at the COMP pin by a factor of 15. This stepped-down voltage is given to one of the comparators as the current reference. The reference to the other comparator, which acts to limit the maximum inductor current, is given externally. It is recommended that the sense resistor RCS be chosen so as to provide about 250mV current sense signal.
Over Voltage Protection
Over voltage protection is achieved by connecting the output voltage to the OVP pin through a resistive divider. The voltage at the OVP pin is constantly compared to the internal 1.25V. When the voltage at this pin exceeds 1.25V, the IC is turned off and FAULT goes low.
Output Short Circuit Protection
The output short circuit condition is indicated by FAULT. At startup, a monoshot circuit, (triggered by the POR circuit), resets an internal flip-flop, which causes FAULT to go high, and remains high during normal operation. This also allows the gate drive to function normally. The steady state current is reflected in the reference voltage connected to the transconductance amplifier. The instantaneous output current is sensed from the FDBK terminal of the amplifier. The short circuit threshold current is internally set to 200% of the steady state current. During short circuit condition, when the current exceeds the internally set threshold, the SR flip-flop is set and FAULT goes low. At the same time, the gate driver of the power FET is inhibited, providing a latching protection. The system can be reset by cycling the input voltage to the IC. Note: The short circuit FET should be connected before the current sense resistor as reversing RS and Q2 will affect the accuracy of the output current (due to the additional voltage drop across Q2 which will be sensed).
Current Limit
Current limit has to be set by a resistor divider from the 1.25V reference available on the IC. Assuming a maximum operating inductor current Ipk (including the ripple current), the voltage at the CLIM pin can be set as: 5 * RSC VCLIM 1.2 * IPK * RCS + * 0.9 RSLOPE Note that this equation assumes a current limit at 120% of the maximum input current. Also, if VCLIM is greater than 450mV, the saturation of the internal opamp will determine the limit on the input current rather than the CLIM pin. In such a case, the sense resistor RCS should be reduced till VCLIM reduces below 450mV. It is recommended that no capacitor be connected between CLIM and GND. If necessary, the capacitor value must be chosen to be less than 1000pF.
Synchronization
The SYNC pin is an input/output (I/O) port to a fault tolerant peer-to-peer and/or master clock synchronization circuit. For synchronization, the SYNC pins of multiple HV9911 based converters can be connected together, and may also be connected to the open drain output of a master clock. When connected in this manner, the oscillators will lock to the device with the highest operating frequency. When synchronizing multiple ICs, it is recommended that the same
11
FAULT Protection
The HV9911 has built-in output over-voltage protection and output short circuit protection. Both protection features are latched, which means that the power to the IC must be recycled to reset the IC. The IC also includes a FAULT
HV9911
timing resistor, corresponding to the switching frequency, be used in all the HV9911 circuits. In rare occasions, given the length of the connecting lines for the SYNC pins, a resistor between SYNC and GND may be required to damp any ringing due to parasitic capacitances. It is recommended that the resistor chosen be greater than 300k. When synchronized in this manner, a permanent HIGH or LOW condition on the SYNC pin will result in a loss of synchronization, but the HV9911 based converters will continue to operate at their individually set operating frequency. Since loss of synchronization will not result in total system failure, the SYNC pin is considered fault tolerant. Note: The HV9911 is designed to SYNC up to four ICs at a time without the use of an external buffer. To SYNC more than four ICs, it is recommended that a buffered external clock be used. recommended that the PWMD pin be used to get zero output current rather than pull the IREF pin to GND.
PWM Dimming
PWM dimming can be achieved by driving the PWMD pin with a TTL compatible source. The PWM signal is connected internally to the three different nodes - the transconductance amplifier, the FAULT output, and the GATE output. When the PWMD signal is high, the GATE and FAULT pins are enabled, and the output of the transconductance opamp is connected to the external compensation network. Thus, the internal amplifier controls the output current. When the PWMD signal goes low, the output of the transconductance amplifier is disconnected from the compensation network. Thus, the integrating capacitor maintains the voltage across it. The GATE is disabled, so the converter stops switching and the FAULT pin goes low, turning off the disconnect switch. The output capacitor of the converter determines the PWM dimming response of the converter, since it has to get charged and discharged whenever the PWMD signal goes high or low. In the case of a buck converter, since the inductor current is continuous, a very small capacitor is used across the LEDs. This minimizes the effect of the capacitor on the PWM dimming response of the converter. However, in the case of a boost converter, the output current is discontinuous, and a very large output capacitor is required to reduce the ripple in the LED current. Thus, this capacitor will have a significant impact on the PWM dimming response. By turning off the disconnect switch when PWMD goes low, the output capacitor is prevented from being discharged, and thus the PWM dimming response of the boost converter improves dramatically. Note: Disconnecting the capacitor might cause a sudden spike in the capacitor voltage as the energy in the inductor is dumped into the capacitor. This might trigger the OVP comparator if the OVP point is set too close to the maximum operating voltage. Thus, either the capacitor has to sized slightly larger or the OVP set point has to be increased. Note: The HV9911 IC might latch-up if the PWMD pin is pulled 0.3V below GND, causing failure of the part. This abnormal condition can happen if there is a long cable between the PWM signal and the PWMD pin of the IC. It is recommended that a 1k resistor be connected between the PWMD pin and the PWM signal input to the HV9911. This resistor, when placed close to the IC, will damp out any ringing that might cause the voltage at the PWMD pin to go below GND.
Internal 1MHz Transconductance Amplifier
HV9911 includes a built in 1MHz transconductance amplifier, with tri-state output, which can be used to close the feedback loop. The output current sense signal is connected to the FDBK pin and the current reference is connected to the IREF pin. The output of the opamp is controlled by the signal applied to the PWMD pin. When PWMD is high, the output of the opamp is connected to the COMP pin. When PWMD is low, the output is left open. This enables the integrating capacitor to hold the charge when the PWMD signal has turned off the gate drive. When the IC is enabled, the voltage on the integrating capacitor will force the converter into steady state almost instantaneously. The output of the opamp is buffered and connected to the current sense comparator using a 15:1 divider. The buffer helps to prevent the integrator capacitor from discharging during the PWM dimming state.
Linear Dimming
Linear dimming can be accomplished by varying the voltage at the IREF pin, as the output current is proportional to the voltage at the IREF pin. This can be done either by using a potentiometer from the REF pin or by applying an external voltage source at the IREF pin. Note: Due to the offset voltage of the transconductance opamp, pulling the IREF pin very close to GND will cause the internal short circuit comparator to trigger and shut down the IC. This limits the linear dimming range of the IC. However, a 1:10 linear dimming range can be easily obtained. It is
12
HV9911
Avoiding False Shutdowns of the HV9911
The HV9911 has two fault modes which trigger a latched protection mode - an over current (or short circuit) protection, and an over voltage protection. To prevent false triggering due to the tripping of the over voltage comparator, (due to noise in the GND traces on the PCB), it is recommended that a 1nF - 10nF capacitor be connected between the OVP pin and GND. Although this capacitor will slow down the response of the over voltage protection circuitry somewhat, it will not affect the overall performance of the converter, as the large output capacitance in the boost design will limit the rate of rise of the output voltage. In some cases, the over current protection may be triggered during PWM dimming, when the FAULT goes high and the disconnect switch is turned on. This triggering of the over current protection is related to the parasitic capacitance of the LED string (shown as a lumped capacitance CLED in Fig. 4). During normal PWM dimming operation, the HV9911 maintains the voltage across the output capacitor (CO), by turning off the disconnect switch and preserving the charge in the output capacitance when the PWM dimming signal is low. At the same time, the voltage at the drain of the disconnect FET is some non-zero value VD. When the PWM dimming signal goes high, FET Q2 is turned ON. This causes the voltage at the drain of the FET (VD) to instantly go to zero. Assuming a constant output voltage VO, d ( Vo - Vd ) iSENSE = CLED * dt dV = - CLED * d dt In this case, the rate of fall of the drain voltage of the disconnect FET is a large value (since the FET turns on very quickly) and this causes a spike of current through the sense resistor, which could trigger the over current protection (depending on the parasitic capacitance of the LED string). To prevent this condition, a simple RC low pass filter network can be added as shown in Fig. 5. Typical values are RF = 1k and CF = 470pF. This filter will block the FDBK pin from seeing the turn-on spike and normalize the PWM dimming operation of the HV9911 boost converter. This will have minimal effect on the stability of the loop but will increase the response time to an output short. If the increase in the response time is large, it might damage the output current sense resistor due to exceeding its peak-current rating. The increase in the short circuit response time can be computed using the various component values of the boost converter. Consider a boost converter with a nominal output current IO = 350mA, an output sense resistor RS = 1.24W, LED string voltage VO = 100V and an output capacitor CO = 2mF. The disconnect FET is a TN2510N8 from Supertex which has a saturation current ISAT = 3A (at VGS = 6V). The increase in the short circuit response time due to the RC filter can then be computed as: Io t RF * CF * ln 1 ISAT - Io 0.35 A = 1k * 470pF * ln 1 3 A - 0.35 A 66ns This increase is found to be negligible (note that the equation is valid for T << RS * CO. In this case, RS * CO = 2.48s, and the condition holds.
Sizing the Output Sense Resistor
To avoid exceeding the peak-current rating of the output sense resistor during short circuit conditions, the power rating of the resistor has to be chosen properly. In this case, the maximum power dissipated in the sense resistor is: 2 PSC = ISAT * RS = 11W From the datasheet for a 1.24W, 1/4W resistor, the maximum power it can dissipate for a single 1ms pulse of current is 11W. Since the total short circuit time is about 350ns (including the 300ns time for turn off), the resistor should be able to handle the current.
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HV9911
VO CO CLED ROVP2 VD
FAULT
VO CO ROVP2 CLED
VD Q2
FAULT FDBK
Q2 Cf RS iSENSE
FDBK
RS
iSENSE
Fig. 4. Output of the boost converter showing LED parsed capacitance
Fig. 5. Adding a low-pass filter to prevent palse triggering.
16-Lead SOIC Package Outline (NG)
9.9 0.10 16
Note 2 3.90 0.10
6.0 0.20
1
Top View
5O - 15O (4 PLCS)
0.25 - 0.50 Note 3
0O - 8O
0.17 - 0.25 1.75 MAX 1.25MIN 1.27BSC 0.31 - 0.51 0.10 - 0.25 0.40 -1.27
45
Side View
End View
Notes: 1. All dimensions in millimeters; angles in degrees 2. Pin 1 identifier must be located within the indicated area 3. Corner shape may differ from drawing
(The package drawing(s) in this data sheet may not reflect the most current specifications. For the latest package outline information go to http://www.supertex.com/packaging.html.)
Doc.# DSFP-HV9911 NR111406
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